Electronic signal processor

ABSTRACT

An electronic signal processor for processing signals includes a complex first filter, one or more gain stages and a second filter. The first filter is characterized by a frequency response curve that includes multiple corner frequencies, with some corner frequencies being user selectable. The first filter also has at least two user-preset gain levels which may be alternately selected by a switch. Lower frequency signals are processed by the first filter with at least 12 db/octave slope, and preferably with 18 db/octave slope to minimize intermodulation distortion products by subsequent amplification in the gain stages. A second filter provides further filtering and amplitude control. The signal processor is particularly suited for processing audio frequency signals.

RELATED APPLICATIONS

This application is a continuation of application Ser. No. 15/438,354filed Feb. 21, 2017, entitled “Electronic Signal Processor”, which is acontinuation of application Ser. No. 14/974,749 filed Dec. 18, 2015, nowU.S. Pat. No. 9,595,249 entitled “Electronic Signal Processor” which isa continuation of application Ser. No. 14/301,004 filed Jun. 10, 2014now U.S. Pat. No. 9,251,775 entitled “Electronic Signal Processor” whichis a continuation of application Ser. No. 13/331,914 filed Dec. 20, 2011now U.S. Pat. No. 8,779,274 entitled “Electronic Signal Processor” whichis a continuation of application Ser. No. 12/126,460 filed May 23, 2008now U.S. Pat. No. 8,084,679 entitled “Electronic Signal Processor” whichis a continuation of application Ser. No. 10/623,433 filed Jul. 18,2003, entitled “Electronic Signal Processor”, now U.S. Pat. No.7,390,960, all of which are hereby incorporated by reference in theirentirety.

FIELD OF THE INVENTION

The present invention relates generally to electronic signal processors.More particularly, a preferred embodiment of the invention relates toaltering or controlling the tonal qualities of electronic signals, suchas audio signals, and related methods.

BACKGROUND OF THE INVENTION

Various prior art devices exist for modifying the tonal qualities ofelectronic signals. In audio frequency applications, the types ofsignals processed can be speech, musical instruments, synthesizedwaveforms, and the like. Prior art devices for processing musicalinstrument signals generally have a very limited ability to provide themusician with a variety of tonal qualities in the resulting sound. Forexample, prior art circuits exist for processing electric guitar signalsthat have a singular tonal quality, or “sound”. This is a seriouslimitation, since the guitarist must frequently employ a plurality ofdifferent circuits if different “sounds” are desired.

Some schemes exist in the art that include circuits with more than asingular sound. Usually this involves adding additional active circuitsthat the guitarist can activate, as desired. While such an arrangementcan be successful, it also results in much greater total component countand added expense.

In addition, in some applications, it is desirable to deliberately adddistortion to the sound to affect the tonal qualities. For example,deliberately adding distortion to the sound of an electric guitar beganin the 1950's when rock music was becoming popular. At this time, theonly techniques that an electric guitarist has to increase the amount ofdistortion into his sound was to increase the volume of a vacuum tubeamplifier by (1) picking the strings of the guitar harder, (2) turningthe volume of the guitar higher, or (3) turning the volume of theamplifier up; or some combination or variation of all three techniques.However, these techniques have the drawbacks that the guitarist usuallycould still not achieve the desired level of distortion, and/or highsound pressure levels were created that many people find uncomfortableor even distressing.

During the 1960's, the characteristic sound of an overdriven vacuum tubeamplifier was realized while playing at lower volumes by using new typesof circuits. These new circuits were frequently called “fuzzboxes” andwere separate boxes that were external to the amplifier. Fuzzboxestypically employed a cascade or series connection of two or moretransistor amplifier gain stages that had high input-to-output gain andthat were easily overdriven by the output signal from the guitar. Thisprovided a favorable increase in distortion and sustain to the guitarsound. However, it also introduced a new quality to the sound that isdisliked by many guitarists. This quality is often referred to as the“solid-state sound” or the “transistor sound”. Either of these terms hasacquired a very negative connotation to many guitarists. That is, thesolid-state or transistor sound is quite different than the “tubesound”, which was developed by the overdriven vacuum tube amplifiers.

Many guitarists continue to believe that the best distortion sounds comefrom amplifiers that employ tube circuits. While the best solid-stateamplifiers come close, they are frequently considered to be inferior tothe tube amplifiers. Despite the many solid-state amplifiers that havebeen developed and introduced to the marketplace since the 1960's, thesolid-state sound is still not on par with that of the tube amplifiers.Indeed, many different schools of thought exist on why there aredifferences in the sound and feel between the solid-state and tubeamplifiers. Recent attempts to emulate the sound and feel of tubeamplifiers have stagnated.

It has been an objective in the guitar industry for many years todevelop solid-state amplifiers that have the sound and feel of theoverdriven tube amplifier. “Feel” indicates that a tube amplifier alsohas a certain tactile quality when overdriven. Many guitarists thinkthat the tube amplifiers respond to the guitarists “touch”, includingtheir picking techniques and playing style, better than the solid-stateamplifiers. In this respect, it is frequently stated that tubeamplifiers are very touch sensitive.

There has been a long-felt need for a solid-state amplifier or signalprocessor that emulates the sound and feel of an overdriven vacuum tubeamplifier.

A need also exists for a signal processor that emulates the sound of anoverdriven vacuum tube amplifier in which the tone may be adjusted orcustomized to the user's desires.

Accordingly, it is a general object of the present invention to providea new and improved signal processor that emulates the sound and feel ofan overdriven vacuum tube amplifier.

Another object of the present invention is to provide a signal processorof the solid-state type that emulates the desired performancecharacteristics of a tube amplifier.

Yet another object of the present invention is to provide a signalprocessor with sound characteristics that may be adjusted to the user'stastes.

A further object of the present invention is to filter the lowerfrequency input signals with a second order or third order high passfilter before amplification of the input signals to reduce lowerfrequency intermodulation distortion when the amplifier is overdriven.

A still further object of the present invention is to provide at leasttwo individual gain controls with overlapping gain characteristics thatmay be switched to provide selectable gain of those frequencies in thepassband of the input filter.

Another object of the present invention is to provide related methods offiltering an input signal with an input filter of the second or thirdorder high pass type to substantially reduce lower frequencyintermodulation distortion in the signal processor.

BRIEF SUMMARY OF THE INVENTION

This invention is directed to an electronic signal processor that hasimproved ability to alter the tonal characteristics of an audiofrequency input signal and to reduce lower frequency intermodulationdistortion. The signal processor may have a buffer stage to receive theinput signal and to provide an input signal with low output impedance tothe first filter of the signal processor.

A first filter is preferably a second or third order high pass filterwith a frequency response curve of 12 db/octave slope or 18 db/octaveslope for the lower frequencies, respectively. One of the purposes ofthe first filter is to substantially reduce lower frequencyintermodulation distortion by means of such filtering. The first filteralso has at least some user-selectable corner frequencies in itsfrequency response curve so that the user may customize the tonalquality of the signal processor. The first filter preferably alsoincludes at least two adjustable gain levels with overlapping gaincharacteristics that may be pre-set by the user and that may bealternately selected. The multiple, user-preset, selectable gain levelsallow the user to adjust the amount of distortion present in, andtherefore the tonal color of, the processor output.

The output of the first filter is input to one or more limiting gainstages, which are in series or cascade configuration. These gain stagescan increase the amount of distortion present in the processor output.Oppositely poled diodes in the feedback circuits of the amplifiers inthe gain stages limit the output amplitude of the amplifiers andcontribute to the distortion characteristics of the signal processor.Preferably, the gain stages have an additional or second feedbackcircuit that introduces a controlled amount of hysteresis, a nonlineardistortion, in the amplification characteristic of the gain stages.Thus, when the gain stages are overdriven by the input signal, theclipping or distortion in the output signal of the gain stages will beenhanced.

The present invention also relates to amplifiers with two feedback loopsfor use in the gain stages of signal processors. The first feedback loopincludes a resistor, a capacitor and at least two diodes, with thediodes oppositely poled between the output of the amplifier and itsinverting input. The second feedback circuit includes at least oneresistor and at least one capacitor coupled between the output of theamplifier and the input of the gain stage. A resistor preferably couplesthe second feedback loop to the inverting input of the amplifier. Thetwo feedback loops interact to enhance the distortion when the amplifieris overdriven by an input signal.

The output from the gain stages is input to a second filter, which is ofthe low pass type and preferably of the second order low pass type. Theoutput the second filter is provided as the output of the signalprocessor.

Related methods of processing an input signal that includes a band offrequencies to reduce lower frequency intermodulation distortionincludes filtering the input signal with the first filter of the secondor third order type, supplying the filtered signal to the gain stages,amplifying the filtered signal in the gain stages, supplying theamplified signal to a second filter of the low pass type, filtering theamplified signal in the second filter, and supplying the signal from thesecond filter as the output signal of the signal processor. The methodsalso include changing at least some of the corner frequencies in thefrequency response curve of the first filter to change or customize thefrequency response of the first filter. The methods further includeselecting one of the two gain controls in the first filter.

BRIEF DESCRIPTION OF THE DRAWINGS

The features of the present invention which are believed to be novel areset forth with particularity in the appended claims. The invention,together with the further objects and advantages thereof, may best beunderstood by reference to the following description taken inconjunction with the accompanying drawings, in the several figures inwhich like reference numerals identify like elements, and in which:

FIG. 1 is a block diagram of the signal processor of the presentinvention;

FIG. 2 is a schematic circuit diagram of a preferred embodiment of thesignal processor of the present invention;

FIG. 3 is a schematic circuit diagram of a preferred embodiment of aninput filter for the signal processor shown in FIGS. 1 and 2;

FIG. 4 is a frequency response curve of the input filter of the blockdiagram shown in the schematic circuit diagram of FIG. 3 under selectedcircuit conditions;

FIG. 5 is a frequency response curve of the input filter shown in theschematic circuit diagram of FIG. 3 under selected circuit conditions;

FIG. 6 is a frequency response curve of the input filter shown in theschematic circuit diagram of FIG. 3 under selected circuit conditions;

FIG. 7 is a frequency response curve of the input filter shown in theschematic circuit diagram of FIG. 3 under selected circuit conditions;

FIG. 8 is a schematic circuit diagram of a preferred embodiment of anamplifier stage for the signal processor shown in FIG.2;

FIG. 9 is a schematic circuit diagram of an alternate embodiment of anamplifier stage for the signal processor shown in FIG. 2;

FIG. 10 is a frequency response curve of the amplifier stages shown inthe schematic circuit diagrams of FIGS. 8 and 9;

FIG. 11 is a schematic circuit diagram of an output filter for thesignal processor shown in FIG. 1;

FIG. 12 is a frequency response curve of the output filter shown in theschematic circuit diagram of FIG. 11 under selected circuit conditions;

FIG. 13 is a block diagram that is related to the block diagram of FIG.1, but with the preferred frequency responses of the first and secondfilters inserted in the respective filter blocks;

FIG. 14 is an alternate embodiment of the frequency response curve forthe first filter k1 shown in the block diagram of FIG. 1; and

FIG. 15 is an alternate embodiment of the frequency response curve forthe second filter k2 shown in the block diagram of FIG. 1.

DETAILED DESCRIPTION OF THE INVENTION

The present invention of a signal processing circuit, generallydesignated 40, is shown in block diagram format in FIG. 1. An inputsignal is received at an input terminal 41 to a small magnitude outputimpedance stage 43. Stage 43 preferably has an output impedance that issignificantly smaller than the input impedance of a first filter k1 44so as not to materially affect the corner frequencies of the firstfilter 44. First filter 44 is a complex filter with multipleuser-adjustable corner frequencies and passband gains. The output offilter 44 is input into a first gain stage 45. The output of the firstgain stage 45 is input into a second gain stage 46. The output of thesecond gain stage 46 is input into a second filter k2 47, which providesthe output signal of the signal processing circuit 40 at a terminal 42.

A preferred schematic for the signal processor circuit 40 is shown inFIG. 2 with the blocks identified in FIG. 1 shown in dashed lines aboutcertain components of the schematic diagram. The design and operation ofcircuit 40 will now be further considered in its various portionscorresponding to the blocks 43-47 shown in FIGS. 1-2.

In general an input signal, such as from a guitar, is buffered by thelow output impedance stage 43 before presentation to the first filter44. For example, as shown in FIG. 2, the low output impedance stage 43may consist of an amplifier 50 that is configured for unity gain. Whilenot shown in block 43 of FIG. 2, it may also be desirable to provide lowpass filtering at the input terminal 41. For example, frequencies abovethe audio band, such as radio frequency interference (RFI) or the like,may be attenuated at or near the input to amplifier 50.

First filter 44 shown in FIG. 3 provides filtering of the lowfrequencies in the audio frequency range to prevent the generation ofsignificant amounts of low frequency intermodulation (IMD) signals,which may result from the subsequent amplification by the first andsecond gain stages 45 and 46. First filter 44 receives its input signalfrom the output of the low impedance stage 43 at an input terminal 51. Aresistor 52 and a capacitor 53, connected in series, receive signalspresent on input terminal 51. An opposite terminal of capacitor 53 isreferenced to ground by a resistor 54.

A single pole, multiple throw switch 55, which may be a rotary switchwith n positions, is connected to capacitor 53 and resistor 54. Switch55 selects one of n capacitors, such as capacitors 56-63 in the exampleshown in FIG. 5. Opposite ends of capacitors 56-63 are connected to acommon node 65.

A double pole, double throw switch 75 selects one of two networks thatare also connected to node 65. In the position shown in FIG. 3, switch75 selects the first network that includes a pair of resistors 66 and68. Resistor 68 may be in the form of an adjustable resistor orpotentiometer with an adjustable terminal 67 to control the amplitude ofthe signals provided through filter 44. If switch 75 is in the oppositeposition from that shown in FIG. 3, the second network consisting ofresistor 70, capacitor 69 and variable resistor or potentiometer 73 isselected. This second network also provides control of the amplitude ofthe signals provided through filter 44 by varying the position of theadjustable terminal 72 of variable resistor 73. In addition, capacitor69 provides some additional filter effects over that of the firstnetwork consisting of resistors 66 and 68.

Whichever network is selected by switch 75 provides the signals thoughthe series connection of a capacitor 76 and a resistor 77 to theinverting input of an operational amplifier 80. Op amp 80 has itsnon-inverting terminal referenced to ground. Op amp 80 also has a pairof diodes 81 and 82 oppositely poled between the output terminal and theinverting terminal of op amp 80 to keep op amp 80 from being overdriven.A resistor 84 and a capacitor 83 are also connected as feedbackcomponents, in parallel with diodes 81-82, between the output terminaland inverting terminal of op amp 80. Op amp 80 also provides the outputsignal of first filter 44 at an output terminal 85.

First filter 44 provides different rates of signal gain or attenuationover different frequency ranges. In the illustrated embodiment of firstfilter 44, there are four corner frequencies f1, f2, f3 and f4, whereeach corner frequency is defined by the known equation f=1/(2πRC) andwhere R is the effective resistance at the frequency of interest, C isthe effective capacitance at the frequency of interest and 7C is thewell-known value of 3.1415 . . . .

FIGS. 4 through 7 illustrate the different effects that are provided bythe first filter 44. While FIGS. 4-7, 10 and 12 do not have a scalealong the frequency axis, it will be understood that these frequencyresponse charts generally cover the frequency range of about 0 Hz to 20KHz, which includes the audio frequency range, which is often specifiedas 20 Hz to 20 KHz. As will be presented more fully below, the frequencyresponse of the first filter 44 depends upon which of capacitors 56-63is selected by switch 55, the first or second network selected by switch75, and the position or adjustment selected for potentiometers 68 or 73.Irrespective of these selections, the gain versus frequency graphs shownin FIGS. 4-7 will, in general, have a slope of 18 db/octave in a firstfrequency band from 0 Hz to f1, 12 db/octave in a second frequency bandfrom f1 to f2, 6 db/octave in a third frequency band from f2 to f3, 0db/octave in a fourth frequency band (which may also be referred to as apassband) from f3 to f4, and −6 db/octave for frequencies above f4.

Filters, such as the first filter 44 that exhibits a slope of 18db/octave in the lower frequency ranges and a passband of 0 db/octave inthe higher frequency ranges are also known in the art as third orderhigh pass filters. In the example of FIG. 6, there is additionally ahigh frequency rolloff of −6 db/octave above the corner frequency f4.Thus, a filter with the frequency response curve shown in FIG. 4 couldalso be referred to as a third order high pass filter with highfrequency rolloff.

FIG. 4 illustrates the effects of varying the passband gain withpotentiometers 68 or 73, depending upon which of the networks isselected by switch 75. In frequency response graph 130, the gain is sethigher than in the graph 131. Of course, if potentiometer 68 is set atfor a higher gain value than potentiometer 73, the user may switch fromhigher to lower gain (and, hence, from higher to lower volume) bychanging switch 75 from the position shown in FIG. 3 to the oppositeposition, and vice versa. To this end, switch 75 may be a foot-operatedswitch. As illustrated in FIG. 4, the changes in gain tend to havegreater affect on those frequency bands that are less attenuated, suchas those frequencies that lie between f2 to beyond f4. If either ofpotentiometers 68, 73 are adjusted by moving the adjustable terminal 67or 72 to its lower most position, the signal will be completelyattenuated since lower pole of switch 75 is referenced to ground. Thus,potentiometers 68, 73 provide a broad range of signal attenuation.

FIG. 5 illustrates the ability to change the gain characteristics ofthose portions of the frequency response curve below frequency f3,including the frequency of the corner frequency f3. This is accomplishedby changing the position of switch 55 to select one of capacitors 56-63.Capacitors 56-63 are selected to be of different capacitive values toprovide different frequency response characteristics. FIG. 5 shows threedifferent frequency response graphs 132-134 for three differentcapacitive values. Of course, with n capacitors of different capacitivevalue, n different frequency response curves will result instead of thethree shown in FIG. 5. Note also that changing the capacitive value withswitch 55 will also affect the corner frequency f3. In the exampleshown, corner frequency f3 a is associated with frequency response curve132, corner frequency f3 b is associated with frequency response curve133 and corner frequency f3 c is associated with frequency responsecurve 134. In general, a lower capacitive value for one of thecapacitors 56-63 will cause the corner frequencies f1, f2 and f3 toshift toward higher frequencies. For example, in order to provide arange of effects through the selection of one of the n capacitors withswitch 55 for audio signal applications, the capacitor with the lowestvalue preferably moves the 12 db/octave slope up to about 4 to 5 KHz. Onthe other hand, the capacitor with the highest capacitive value selectedby switch 55 preferably moves the 12 db/octave slope down to about 30Hz. Thus, the lower frequencies that the 12 db/octave portion of thefrequency response curve operates on can range from about 30 Hz to about5 KHz. The actual selection will depend upon the preferences of theuser.

FIG. 6 illustrates the ability to change the gain characteristics ofthat portion of the frequency response curve above the corner frequencyf4. The feedback components, capacitor 83 and resistor 84, across op amp80 normally determine the frequency of corner frequency f4 a when switch75 is in the position shown in FIG. 3. This results in the frequencyresponse graph shown by graph 136. However, when switch 75 is in theopposite position to that shown in FIG. 3, capacitor 69 will change thefrequency response to a graph such as graph 135 in FIG. 6. Note that ingraph 135, capacitor 69 also causes an increase in the corner frequencyf4 b above that of f4 a, and an increase in the higher frequency gainabove that of graph 136.

FIG. 7 is a composite of the frequency response graphs of FIGS. 4-6. Thefrequency shifts of some of the corner frequencies have not beenillustrated, as in FIGS. 4-6, for purposes of simplifying this compositegraph. It will thus be appreciated that the above-described differingtechniques for customizing the frequency response characteristics of thefirst filter 44 provide the ability to customize or fine tune anyportion of the audio frequency spectrum, as desired by the user.

The preferred embodiment of an amplifier for the first gain stage 45 inFIG. 3 is shown in FIG. 8. An input terminal 88 of the first gain stage45 passes input signals through a resistor 89 and a capacitor 90 to anode 97. Node 97 is connected via a feedback resistor 91 to the outputterminal of an op amp 98 and via a resistor 96 to the inverting input ofop amp 98. The non-inverting input of op amp 98 is referenced to ground.Feedback components, including a capacitor 94 and a resistor 95, areconnected from the inverting input to the output of op amp 98.Oppositely poled diodes 92 and 93, also connected from the invertinginput to the output of op amp 98, keep the op amp output amplitudelimited. Diodes 92-93 clip symmetrically and therefore tend to limit theamount of distortion when the op amp 98 is overdriven. Diodes 92-93 alsotend to provide some nonlinear distortion such as hysteresis when op amp98 is overdriven since the feedback capacitor 94 will be charged byconduction of diodes 92-93. However, when diodes 92-93 becomenon-conductive, the impedance seen by feedback capacitor 94 increasesand capacitor 94 takes longer to discharge. Thus, the first feedbackcircuit consisting of diodes 92-93, capacitor 94 and resistor 95operates in two different impedance modes, depending upon whether diodes92-93 are conductive or non-conductive.

The amplifier embodiment of FIG. 8 has superior performancecharacteristics when used in signal processors for guitars. It isdesirable for the best tonal characteristics resulting from clippingcaused by gain stage 45, when overdriven, that the clipping not besymmetrical. To this end, a second feedback circuit, consisting ofresistors 89 and 91 and capacitor 90, creates additional nonlineardistortion such as hysteresis in the response of the gain stage 45.Resistor 96 provides some interaction between the first feedback circuitconsisting of resistor 95, capacitor 94 and diodes 92-93, and the secondfeedback circuit. This additional nonlinear distortion such ashysteresis provides further distortion of the input signal by gain stage45 when the op amp 98 is overdriven.

A simplified gain stage, generally designated 48, is shown in FIG. 9,may be used in place of the gain stage 45 of FIG. 8, if desired.Simplified gain stage 48 is similar in structure and operation to gainstage 45, except that resistors 91 and 96 of gain stage 45 that form aportion of an additional feedback loop about op amp 98 in FIG. 8 areeliminated. Thus, the operation of gain stage 48 is similar in operationto the op amp 80 in the first filter 44, as described above.

The gain stages employed in the second gain stage 46 in FIG. 1 arepreferably similar to those used in the first gain stage, and as shownin FIG. 8 or FIG. 9. However, the second gain stage may have pairs ofdiodes 104-105 and 106-107 oppositely poled across the op amp 112 asshown in the complete schematic of FIG. 2 to allow for greater amplitudesignals before the diodes 104-107 become operative and limit the outputamplitude.

Second gain stage 46 is connected in series or cascade with the firstgain stage 45. Each of gain stages 45, 46 preferably has a gain ofgreater than one and is nominally inverting. The frequency response forgain stages 45 or 46 is shown by a graph 137 in FIG. 10, and has a lowercorner frequency f1 and a higher corner frequency fh. From 0 Hz to f1,the slope is 6 db/octave. From f1 to fh, which is the passband, theslope is 0 db/octave. At frequencies above fh, the slope is −6db/octave.

The second filter stage, generally designated 47, is shown in FIG. 11.An input terminal 116 receives input signals from the output terminal ofthe second gain stage 46. Input terminal 116 is connected via a resistor117 and capacitor 118 to a node 122. A resistor 119 and a capacitor 120are connected in series between node 122 and ground. Node 122 is alsoconnected via a resistor 121 to another node 127. A resistor 123 and acapacitor 124 are connected in series between node 127 and ground. Alsoseparately connected in parallel between node 127 and ground are acapacitor 125 and a potentiometer 126. The variable wiper arm ofpotentiometer 126 is connected to the output terminal 42 of the signalprocessor 40 of FIG. 2. Potentiometer 126 may function as the volumecontrol for the signal processor.

The second filter 47 may have a complex frequency response as shown bythe graph 138 in FIG. 12. Graph 138 may have six positive cornerfrequencies, f5, f6, f7, f8, f9 and f10, in order of increasingfrequency. From 0 Hz to corner frequency f5, the slope is 6 db/octave;from corner frequency f5 to corner frequency f6, the slope is 0db/octave; from corner frequency f6 to corner frequency f7, the slope is−6 db/octave; from corner frequency f7 to corner frequency f8, the slopeis −12 db/octave; from corner frequency f8 to corner frequency f9, theslope is −6 db/octave; from corner frequency f9 to corner frequency f10,the slope is 0 db/octave; and above corner frequency f10, the slope is−6 db/octave. Capacitor 118 creates the low frequency rolloff belowcorner frequency f5, and capacitor 125 creates the high frequencyrolloff above corner frequency f10.

FIG. 13 is a block diagram that is related to the block diagram shown inFIG. 4, but with the preferred frequency responses of the first andsecond filters 44, 47 shown in the filter blocks. In addition, the twogain stages 45-46 are shown combined in FIG. 13 into a single stage.While preferred embodiments of the circuitry for the filters 44, 47 havebeen presented above in FIGS. 3 and 11, it will be appreciated by thoseskilled in the art that these filters could be active or passive andprovide the desired frequency response curves. In accordance with oneaspect of the present invention, at least 12 db/octave is used in thelower frequencies of the audio spectrum to provide greater attenuationof the lower audio frequencies. This helps minimize the production oflower frequency intermodulation distortion (IMD) frequency products, aspreviously discussed above, by the significant gain of the gain stages45-46. This avoids the commonly known muddy sound produced by prior artamplifiers.

The gain stages 45-46 may be combined into a single gain, or constitutea plurality of individual gain stages coupled together in the knowncascade configuration.

The distortion produced may be modified by providing some offset voltageto the operational amplifiers, such as by referencing the non-invertinginputs to op amps 98 and 112 in FIGS. 2 and 8-9 to a reference (bias)voltage instead of to ground. Such use of bias voltage may be necessaryif the op amps have unequal positive and negative supply voltages. Theseop amps 98 and 112 operate linearly so long as they are not overdriven.As previously discussed, if the op amps 98 and 112 are overdriven, thefeedback diodes 92-93 and 104-107 will be rendered conductive. Thus, inthe preferred embodiment of the invention, non-linearity of the gainstages results when these normally nonconductive diodes becomeconductive. These non-linearities may be modified, if desired, by offsetbiasing of the op amps 98 and 112, such as by biasing the non-invertinginputs at a nonzero reference voltage.

An alternative frequency response curve 141 is shown in FIG. 14 for thefirst filter 44, instead of the frequency responses shown in FIGS. 4-7.In this embodiment, frequency response curve 141 has a slope of 12db/octave at the lowest frequencies instead of 18 db/octave below thecorner frequency f1 in FIGS. 4-7. Curve 141 also does not have the highfrequency rolloff of −6 db/octave for the higher frequencies, such asabove the corner frequency f4 in FIGS. 4-7. Characteristics of curve 141can be provided by eliminating capacitors 53 and 83 in the schematic offilter 44 in FIG. 3. For example, short circuiting of capacitor 53 willeliminate the additional 6 db/octave of slope at the lowest frequenciesof interest, thereby also eliminating the corner frequency f1.Elimination of capacitor 83 will also eliminate the corner frequency f4in FIGS. 4-7 and the −6 db/octave rolloff for frequencies above f4.However, since capacitor 83 also contributes to the stability of op amp80, it may be desirable to simply decrease the capacitive value ofcapacitor 83 such that the corner frequency f4 is above the frequenciesof interest, and which effectively increases the passband of 0 db/octaveslope. A first filter 44 with the frequency response characteristics ofFIG. 14, instead of with the frequency response characteristics of FIGS.4-7, will provide sufficient attenuation of the lower frequencies priorto amplification by the gain stages 45-46 to minimize IMD frequencyproducts in many applications.

An alternative frequency response curve 142 is shown in FIG. 15 for thesecond filter 47, instead of the frequency response curve 138 shown inFIG. 12. In this embodiment, frequency response curve 142 has a slope of0 db/octave at the lowest frequencies instead of 6 db/octave below thecorner frequency f5 in FIG. 12. Curve 142 also does not have the highfrequency rolloff of −6 db/octave for the higher frequencies, such asabove the corner frequency f10 in FIG. 12. A filter having the frequencyresponse curve shown in FIG. 15 is known as a low pass filter. If theslope above the low frequencies is −12 db/octave for n=2, the filter maybe referred to as a second order low pass filter.

The frequency response curve 138 in FIG. 12 may be easily modified toresemble the frequency response curve 142 in FIG. 15 by eliminating thelow frequency rolloff capacitor 118 from the schematic shown in FIG. 11and by eliminating the high frequency rolloff capacitor 125. This willalso eliminate the corner frequencies f5 and f10 shown in FIG. 12.Alternately, capacitor 125 may be decreased in value such that thecorner frequency f10 is moved to a higher frequency beyond the frequencyrange shown in FIG. 12.

While particular embodiments of the invention have been shown anddescribed, it will be obvious to those skilled in the art that changesand modifications may be made therein without departing from theinvention in its broader aspects.

1. An electric filter network system for real-time processing of electrical signals in an electric guitar amplifier, said filter network system including: an input port receiving an input signal derived in real-time from the output of an electric guitar; an output port outputting an output signal derived from said input signal; a first potentiometer, wherein said first potentiometer is adjustable by a user to set at least in part a first voltage gain magnitude value from said input port to said output port at a first frequency; a second potentiometer, wherein said second potentiometer is adjustable by a user to set at least in part a second voltage gain magnitude value from said input port to said output port at said first frequency; a first network including said first potentiometer; a second network including said second potentiometer; a rotary switch, wherein said rotary switch is adapted to select at least in part a capacitance value for said electric filter network system by selecting at least in part a capacitor having said capacitance value from a plurality of capacitors; and a user-operated switch, wherein said user-operated switch is adapted to select at least in part an operating network and an operating voltage gain magnitude value for said electric filter network system by selecting at least in part either said first network and said first voltage gain magnitude value or said second network and said second voltage gain magnitude value, wherein both said first voltage gain magnitude value and said second voltage gain magnitude value are set at least in part by said capacitance value selected at least in part by said rotary switch, wherein said operating network operates in real-time on said input signal to derive at least in part said output signal.
 2. The filter network system of claim 1 further including a first amplifier circuit, wherein said first amplifier circuit is positioned downstream from both said first network and said second network.
 3. The filter network system of claim 2 further including a second amplifier circuit, wherein said second amplifier circuit is positioned downstream from said first amplifier circuit.
 4. The filter network system of claim 3 wherein both said first network and said second network consist of passive circuit components.
 5. The filter network system of claim 1 wherein both of said first potentiometer and said second potentiometer are passive devices and have three terminals.
 6. The filter network system of claim 1 wherein both of said first potentiometer and said second potentiometer are passive rotary potentiometers and have three terminals.
 7. The filter network system of claim 1 wherein said user-operated switch contains only a single switch, wherein said single switch has only a single throw.
 8. The filter network system of claim 1 wherein both said first network and said second network consist of passive circuit components.
 9. An electric filter network system for real-time processing of electrical signals in an electric guitar amplifier, said filter network system including: an input port receiving an input signal derived in real-time from the output of an electric guitar; an output port outputting an output signal derived from said input signal; a first potentiometer, wherein said first potentiometer is adjustable by a user to set at least in part a first voltage gain magnitude value from said input port to said output port at a first frequency; a second potentiometer, wherein said second potentiometer is adjustable by a user to set at least in part a second voltage gain magnitude value from said input port to said output port at said first frequency; a first network including said first potentiometer; a second network including said second potentiometer; a first capacitor, wherein said first capacitor has a first capacitance value, wherein said first capacitor is included in said first network and excluded from said second network, wherein said first capacitance value sets at least in part said first voltage gain magnitude value; a rotary switch, wherein said rotary switch is adapted to select at least in part a second capacitance value for said electric filter network system by selecting at least in part a second capacitor having said second capacitance value from a plurality of capacitors; and a user-operated switch, wherein said user-operated switch is adapted to select at least in part an operating network and an operating voltage gain magnitude value for said electric filter network system by selecting at least in part either said first network and said first voltage gain magnitude value or said second network and said second voltage gain magnitude value, wherein both said first voltage gain magnitude value and said second voltage gain magnitude value are set at least in part by said second capacitance value selected at least in part by said rotary switch, wherein said operating network operates in real-time on said input signal to derive at least in part said output signal.
 10. The filter network system of claim 9 further including a first amplifier circuit, wherein said first amplifier circuit is positioned downstream from both said first network and said second network.
 11. The filter network system of claim 10 further including a second amplifier circuit, wherein said second amplifier circuit is positioned downstream from said first amplifier circuit.
 12. The filter network system of claim 11 wherein both said first network and said second network consist of passive circuit components.
 13. The filter network system of claim 9 wherein both of said first potentiometer and said second potentiometer are passive devices and have three terminals.
 14. The filter network system of claim 9 wherein both of said first potentiometer and said second potentiometer are passive rotary potentiometers and have three terminals.
 15. The filter network system of claim 9 wherein said user-operated switch contains only a single switch, wherein said single switch has only a single throw.
 16. The filter network system of claim 9 wherein both said first network and said second network consist of passive circuit components.
 17. An electric filter network system for real-time processing of electrical signals in an electric guitar amplifier, said filter network system including: an input port receiving an input signal derived in real-time from the output of an electric guitar; an output port outputting an output signal derived from said input signal; a first potentiometer having a first terminal, wherein said first potentiometer is adjustable by a user to set at least in part a first voltage gain magnitude value from said input port to said output port at a first frequency; a second potentiometer, wherein said second potentiometer is adjustable by a user to set at least in part a second voltage gain magnitude value from said input port to said output port at said first frequency; a first network including said first potentiometer; a second network including said second potentiometer; a rotary switch, wherein said rotary switch is adapted to select at least in part a capacitance value for said electric filter network system by selecting at least in part a capacitor having said capacitance value from a plurality of capacitors; a user-operated switch, wherein said user-operated switch is adapted to select at least in part an operating network and an operating voltage gain magnitude value for said electric filter network system by selecting at least in part either said first network and said first voltage gain magnitude value or said second network and said second voltage gain magnitude value, wherein both said first voltage gain magnitude value and said second voltage gain magnitude value are set at least in part by said capacitance value selected at least in part by said rotary switch; and a third potentiometer having a second terminal, wherein said third potentiometer is adjustable by a user to set at least in part both said first voltage gain magnitude value and said second voltage gain magnitude value, wherein said second terminal is positioned downstream from said first terminal, wherein said operating network operates in real-time on said input signal to derive at least in part said output signal.
 18. The filter network system of claim 17 further including a first amplifier circuit, wherein said first amplifier circuit is positioned downstream from both said first network and said second network.
 19. The filter network system of claim 18 further including a second amplifier circuit, wherein said second amplifier circuit is positioned downstream from said first amplifier circuit.
 20. The filter network system of claim 19 wherein both said first network and said second network consist of passive circuit components.
 21. The filter network system of claim 17 wherein both of said first potentiometer and said second potentiometer are passive devices and have three terminals.
 22. The filter network system of claim 17 wherein both of said first potentiometer and said second potentiometer are passive rotary potentiometers and have three terminals.
 23. The filter network system of claim 17 wherein said user-operated switch contains only a single switch, wherein said single switch has only a single throw.
 24. The filter network system of claim 17 wherein both said first network and said second network consist of passive circuit components. 